This invention relates to a switching power supply circuit having a power factor improving function.
Various power supply circuits in the form of a composite resonance type converter wherein a resonance type converter is provided on the primary side and a resonance circuit is provided also on the secondary side have been proposed by the assignee of the present application. Also various power circuits which include a power factor improvement circuit for improving the power factor of a composite resonance type converter have been proposed by the assignee of the present application.
Among such power factor improving circuits, power factor improving circuits of the voltage feedback type have been proposed wherein a voltage resonance pulse voltage generated on the primary side is fed back to a smoothing capacitor to increase the conduction angle of alternating-current input current to improve the power factor. As power factor improving circuits of the type described, various circuits including a circuit of the electrostatic capacity coupling type having a capacitor voltage dividing scheme, a circuit of the magnetic coupling type of a capacity voltage dividing scheme, a circuit of the magnetic coupling type of a tertiary winding scheme and a circuit of the diode coupling type of a tertiary winding scheme have been proposed by the assignee of the present patent application.
It is considered that, among the various power factor improving circuits mentioned, the power factor improving circuit of the diode coupling type of a tertiary winding scheme is most useful in terms of the power conversion efficiency, the cost, the variation characteristic of the direct-current input voltage, the zero volt switching (ZVS) operation region of a switching element and so forth.
Here, an example of a switching power supply circuit which includes a power factor improving circuit of the diode coupling type of a tertiary winding scheme as a related art apparatus is described with reference to FIG. 17.
Referring to FIG. 17, the power supply circuit shown includes a power factor improving circuit 20 for improving the power factor of a switching converter of the voltage resonance type.
The power supply circuit further includes a line filter 21 provided for a commercial alternating-current power AC and formed from, for example, a line filter transformer or an across-the-line capacitor (capacitor for filtering noise of a power supply). The power supply circuit further includes a bridge rectifier circuit Di for full-wave rectifying the commercial alternating-current power AC.
A rectification output of the bridge rectifier circuit Di is charged into a smoothing capacitor Ci through the power factor improving circuit 20. Thus, a rectified and smoothed voltage Ei is obtained across the smoothing capacitor Ci.
The voltage resonance type converter includes a switching element Q1 in the form of, for example, a MOS-FET.
A clamp diode DD is interposed between the drain and the source of the switching element Q1 such that it forms a path for clamp current which flows when the switching element Q1 is off.
The drain of the switching element Q1 is connected to a positive terminal of a smoothing capacitor Ci through a primary winding N1 of an isolating converter transformer PIT. The source of the switching element Q1 is connected to a ground on the primary side.
A switching driving signal from a switching driving circuit not shown is applied to the gate of the switching element Q1 so that the switching element Q1 performs a switching operation in response to the switching driving signal. The switching driving signal has a frequency which is varied, for example, in response to the level of a secondary-side direct-current output voltage. Thus, the secondary-side direct-current output voltage is stabilized through the switching frequency control.
Further, a parallel resonance capacitor Cr is connected between the drain and the source of the switching element Q1. The parallel resonance capacitor Cr has a capacitance which cooperates with a leakage inductance L1 of the primary winding N1 side of the isolating converter transformer PIT to form a primary-side parallel resonant circuit of the voltage resonance type converter. When the switching element Q1 is off, the parallel resonant circuit acts such that the voltage across the resonance capacitor Cr actually exhibits a pulse waveform of a sine wave, whereby operation of the voltage resonance type is obtained.
One end of the primary winding N1 of the isolating converter transformer PIT is connected to the drain of the switching element Q1 while the other end of the primary winding N1 is connected to the positive electrode (rectified and smoothed voltage Ei) of the smoothing capacitor Ci.
A tertiary winding N3 as a separate winding is formed at the same place as that of the primary winding N1, that is, on the primary side. The tertiary winding N3 functions as a feedback winding, and a terminal end of the tertiary winding N3 is connected to an anode point of a high speed recovery type diode D3 of the power factor improving circuit 20 through a series resonant capacitor C10.
On the secondary side of the isolating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in a secondary winding N2. In this instance, since a secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2, a parallel resonant circuit is formed from a leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2. By the parallel resonant circuit, the alternating voltage excited in the secondary winding N2 becomes a resonance voltage. In other words, a voltage resonance operation is obtained on the secondary side.
In particular, in the power supply circuit, a parallel resonant circuit for providing a switching operation of the voltage resonance type is provided on the primary side while another parallel resonant circuit for obtaining a voltage resonance operation is provided on the secondary side. In other words, a composite resonance type switching converter is formed in the power supply circuit.
In this instance, a rectification diode Do1 and a smoothing capacitor Co1 are connected in such a manner as seen in FIG. 17 to the secondary side parallel resonant circuit formed in such a manner as described above such that a half-wave rectifying smoothing circuit, which forms a direct-current output voltage Eo1, is formed.
Now, a configuration of the power factor improving circuit 20 is described.
In the power factor improving circuit 20, a choke coil Ls and the high speed recovery type diode D3 are connected in series and interposed between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci.
A filter capacitor CN is connected in parallel to the series connection of the choke coil Ls and the high speed recovery type diode D3 so as to form a low pass filter of the normal mode together with the choke coil Ls.
The tertiary winding N3 of the isolating converter transformer PIT is connected to a node between the anode of the high speed recovery type diode D3 and choke coil Ls of the power factor improving circuit 20 through the series resonant capacitor C10 so that a switching output voltage (voltage resonance pulse voltage) obtained by the primary-side parallel resonant circuit is fed back to the power factor improving circuit 20.
In this instance, when the absolute voltage of an alternating-current input voltage VAC exhibits a value in the proximity of a peak thereof, the high speed recovery type diode D3 conducts and charging current flows from the alternating-current input power supply AC to the smoothing capacitor Ci through the choke coil Ls and the high speed recovery type diode D3. Simultaneously, a voltage resonance pulse voltage of the tertiary winding N3 is fed back to the series circuit of the series resonant capacitor C10 and the high speed recovery type diode D3 to cause the high speed recovery type diode D3 to effect a switching operation to increase the conduction angle of an alternating-current input current IAC thereby to achieve a power factor improving function.
If the absolute value of the alternating-current input voltage VAC decreases, then the high speed recovery type diode D3 is rendered non-conducting, and the tertiary winding N3 which provides the voltage resonance pulse voltage cooperates with the series circuit of the series resonant capacitor C10, choke coil Ls and filter capacitor CN to form a series resonant circuit.
FIGS. 18 and 19 show operation waveforms of the components of the circuit described above. Particularly, FIG. 18 shows operation waveforms when the alternating-current input voltage VAC has a value around the zero voltage while FIG. 19 shows operation waveforms when the alternating-current input voltage VAC has a value around a peak voltage.
Referring to FIG. 18, it can be seen from the waveforms of the current iQ and the voltage vds of the switching element Q1 that the operation of the circuit described above is a ZVS operation and the switching loss can be reduced.
Further, the circuit generates a voltage of a waveform similar to that of the voltage vds of the switching element Q1 as a tertiary winding voltage V3. When the voltage V3 is applied to the series resonant capacitor C10, choke coil Ls and filter capacitor CN and resonance current flows, the anode terminal voltage of the high speed recovery type diode D3 oscillates in a switching period. When the alternating-current input voltage VAC is around 0, the input rectification voltage V1 is low, and therefore, the anode voltage of the high speed recovery type diode D3 wherein a voltage generated by the choke coil Ls is superposed on the input rectification voltage V1 is normally lower than the cathode voltage which is the voltage Ei across the smoothing capacitor Ci and the high speed recovery type diode D3 remains in an off state. Accordingly, no alternating-current input current flows.
If the alternating-current input voltage VAC rises until it exceeds the input rectification voltage V1, then since the anode voltage of the high speed recovery type diode D3 becomes higher than the input smoothed voltage Ei due to the voltage superposed thereon, the high speed recovery type diode D3 is rendered conducting and the alternating-current input current IAC begins to flow through the high speed recovery type diode D3. Accordingly, since the alternating-current input current TAC begins to flow at a timing of the alternating-current input voltage VAC which is lower by the voltage generated by the choke coil Ls than the input smoothed voltage Ei, the conduction angle of the alternating-current input current IAC increases and the power factor can be improved.
Incidentally, there is a demand to make a switching power supply circuit ready for both of the 100 V type and the 200 V type as the alternating-current input voltage VAC in order to make it possible for the switching power supply circuit to be ready for worldwide use.
In order for the switching power supply circuit to satisfy a requirement for a load power variation from 200 W to 0 W against an input variation between the 100 V type and the 200 V type, an active voltage clamp circuit must be added to the primary side of the composite resonance type converter to increase the control range of the switching operation.
In this instance, it is a possible idea to dispose an active voltage clamp circuit on the primary side of such a circuit as shown in FIG. 17 which includes the power factor improving circuit 20 of the diode coupling-type of a tertiary winding scheme. The circuit arrangement, however, has the following problems.
Even if, where the alternating-current input voltage VAC is of the 100 V type, the power factor is improved to approximately 0.85 and a harmonic distortion control value is satisfied, where the alternating-current input voltage VAC is 230 V, the power factor exhibits a drop to approximately 0.7 and a harmonic distortion control value is not satisfied. Therefore, a power supply circuit of an improved power factor ready for worldwide use cannot be achieved.
Further, the circuit described above exhibits a great drop of the power factor as the load power decreases and does not make a power supply of a stabilized improved power factor which operates suitably in response to a variation of the load.
For example, FIG. 20 illustrates a variation characteristic of the power factor with respect to the load current of the circuit described above with reference to FIG. 17. As seen FIG. 20, in the variation characteristic shown, the power factor drops as the load current decreases.
Meanwhile, in order to assure a ZVS operation region, the series resonance frequency of the power factor improving circuit 20 need be set lower than the switching frequency.
When the alternating-current input voltage VAC is low, if the high speed recovery type diode D3 is ignored because it is off, then the power factor improving circuit 20 is regarded as an LC series resonant circuit wherein the tertiary winding N3 serves as a voltage source. If the switching frequency is lower than the series resonance frequency, then since the LC series resonant circuit acts as a capacitive circuit at the frequency, the current flowing therethrough has a leading phase with respect to the voltage V3 generated in the tertiary winding N3. Since the voltage V3 induced has a waveform similar to the voltage vds across the switching element Q1, at a point of time when the resonance voltage of the switching element Q1 decreases until it reaches a voltage proximate to 0, the current flows from the series resonant capacitor C10 toward the tertiary winding N3. The voltage vds across the switching element Q1 is used to charge or discharge the parallel resonance capacitor Cr through the inductances L1 and L2 until 0 volt is reached to realize a ZVS operation of the switching element Q1. In the case described above, however, since the current which should discharge the parallel resonance capacitor Cr through the inductances L1 and L2 is weakened with the current to be supplied from the tertiary winding N3 to the primary winding N1, the parallel resonance capacitor Cr cannot be discharged fully, which disables the ZVS operation. Consequently, when the switching element Q1 is switched on, switching loss is generated and drops the efficiency.
Consequently, the series resonance frequency of the power factor improving circuit 20 need be set lower than the switching frequency as described above. This, however, provides restriction to the value of the inductance Ls of the power factor improving circuit 20 and the electrostatic capacitance value of the series resonant capacitor C10 and makes optimum designing difficult.